|
|
| United States Patent Application |
20070155348
|
| Kind Code
|
A1
|
|
Razavi; Behzad
; et al.
|
July 5, 2007
|
Apparatus and method for ultra wide band architectures
Abstract
The present invention describes a transmitter/receiver architecture that
uses a Weaver architecture in conjunction with digitally controlled
adder/subtractor components to insert/extract a signal into/from the
multi-channel system. In the transmitter, the selection of the band
select bit causes the up/downconverted IF baseband I and Q signals to
insert/extract on either side of an RF LO signal. In addition, the image
of the first LO is eliminated while the desired signal is enhanced after
passing through this new architecture. The invention also adds an RSSI
circuit to the MBOA Weaver architecture receiver architecture to detect
whether an 802.11 WLAN signal is interfering with the desired UWB signal.
If so, the system is designed to detect this interference and jump to a
new frequency range to avoid this interference. This invention focuses on
devices that operate over the entire UWB band including the newly formed
60 GHz UWB band system.
| Inventors: |
Razavi; Behzad; (Los Angeles, CA)
; Soe; Zaw; (Encinitas, CA)
|
| Correspondence Name and Address:
|
THADDEUS GABARA
62 BURLINGTON ROAD
MURRAY HILL
NJ
07974
US
|
| Assignee Name and Adress: |
WIONICS RESEARCH
Irvine
CA
|
| Serial No.:
|
321348 |
| Series Code:
|
11
|
| Filed:
|
December 29, 2005 |
| U.S. Current Class: |
455/118; 455/112 |
| U.S. Class at Publication: |
455/118; 455/112 |
| Intern'l Class: |
H04B 1/04 20060101 H04B001/04 |
Claims
1. A transmitter architecture comprising; an I input signal; a Q input
signal; a RF output signal; an element comprising; a first mixer coupled
to a first input signal and an I LO signal; a second mixer coupled to a
second input signal and a Q LO signal; an output of the first mixer is
coupled to an adder/subtractor; an output of the second mixer is coupled
to the adder/subtractor; wherein the adder/subtractor combines the output
of the two mixers determined by a digital band select bit; and at least
three elements are coupled together; such that the first element
upconverts the I and Q input signals to generate an IF_I signal; the
second element upconverts the I and Q input signals to generate an IF_Q
signal; and the third element upconverts the IF_I and IF_Q signals to
generate the RF output signal.
2. The transmitter architecture of claim 1, wherein the I input signal
comprises; a baseband component; and the Q input signal comprises; a
baseband component.
3. The transmitter architecture of claim 1, wherein the Q LO signal is in
quadrature to the I LO signal.
4. The transmitter architecture of claim 1, wherein the I and Q LO signals
of the elements have discrete frequency values.
5. The transmitter architecture of claim 1, wherein the I and Q LO signals
of the first and second elements have a frequency different than the I
and Q LO signals of the third element.
6. The transmitter architecture of claim 1, wherein; the I and Q LO
signals of the third element is set to a constant frequency value.
7. The transmitter architecture of claim 1, wherein the IF_I and IF_Q
signals are each coupled to an amplifier.
8. The transmitter architecture of claim 1, wherein a first value of the
digital band select bit subtracts, adds and subtracts the output of the
two mixers in the first, second and third elements, respectively; wherein
a second value of the digital band select bit adds, subtracts and adds
the output of the two mixers in the first, second and third elements,
respectively.
9. The transmitter architecture of claim 1, wherein the selection of the
digital band select bit positions the RF output signal of the upconverted
IF_I and IF_Q signals on either side of the I and Q LO signals of the
third element.
10. The transmitter architecture of claim 1, wherein the elements reside
on an integrated circuit substrate.
11. The transmitter architecture of claim 1, wherein a matching network
couples the RF output signal of the third element to an antenna.
12. The transmitter architecture of claim 1, wherein the RF output signal
is coupled to an antenna.
13. The transmitter architecture of claim 12, wherein the antenna resides
on an integrated circuit substrate.
14. The transmitter architecture of claim 12, wherein the antenna is
formed on a structure independent of the integrated circuit substrate.
15. The transmitter architecture of claim 1, wherein the I and Q input
signals are each coupled to a Low Pass Filter (LPF) before being applied
to the first and second element.
16. The transmitter architecture of claim 15, wherein the I and Q input
signals are each coupled to a Programmable Gain Amplifer (PGA) before
being applied to the LPFs.
17. A transmitter architecture comprising; first means for mixing a first
and a second input signal with a first quadrature LO; means for
generating a first and second IF signals by combining the outputs of the
first mixing means under control of a band select signal; second means
for mixing the first and second IF signals with a second quadrature LO;
means for generating a RF output signal by combining the outputs of the
second mixing means under control of the band select signal means; and
means for propagating the RF signal using an antenna; wherein the RF
output signal can be shifted by a frequency of the first quadrature LO
above or below a frequency of the second quadrature LO under control of
the band select signal means.
18. The transmitter architecture of claim 17, wherein the first and second
quadrature LO signals have discrete frequency values.
19. The transmitter architecture of claim 17, wherein the first quadrature
LO signal has a frequency different than the second quadrature LO.
20. The transmitter architecture of claim 17, wherein the second
quadrature LO signals is set to a constant frequency value.
21. The transmitter architecture of claim 17, wherein the transmitter
architecture resides on an integrated circuit substrate.
22. The transmitter architecture of claim 21, wherein the antenna resides
on the integrated circuit substrate.
23. The transmitter architecture of claim 21, wherein the antenna is
formed on a structure independent of the integrated circuit substrate.
24. A method of changing a band select bit,causing an IF upconverted
baseband I and Q signal to form on either side of an RF sinusodial signal
comprising the steps of; generating a first IF upconverted signal by
mixing a coupled I baseband signal with an IF I sinusoidal signal;
generating a second IF upconverted signal by mixing a coupled Q baseband
signal with an IF Q sinusoidal signal; generating a third IF upconverted
signal by mixing the coupled I baseband signal with the IF Q sinusoidal
signal; generating a fourth IF upconverted signal by mixing the coupled Q
baseband signal with the IF I sinusoidal signal; coupling the first IF
and the second IF upconverted signals to a first adder/subtractor unit
controlled by the band select bit to generate an IF_I signal; coupling
the third IF and the fourth IF upconverted signals to a second
adder/subtractor unit controlled by the band select bit to generate an
IF_Q signal; generating a first RF upconverted signal by mixing the IF_I
signal with an RF I sinusoidal signal; generating a second RF upconverted
signal by mixing the IF_Q signal with an RF Q sinusoidal signal; and
coupling the first RF and the second RF upconverted signal to a third
adder/subtractor unit controlled by the band select bit to generate an RF
output signal; whereby changing the band select bit causes the IF
upconverted baseband I and Q signal to form on either side of the RF
sinusoidal signal.
25. The method of claim 24, further comprising the steps of amplifying the
coupled I and Q baseband signals; and low pass filtering the coupled I
and Q baseband signals.
26. The method of claim 24, further comprising the step of maintaining the
frequency of the RF I and Q sinusoidal signals constant.
27. The method of claim 24, further comprising the step of amplifying both
of the IF_I and IF_Q signals prior to RF mixing.
28. The method of claim 24, further comprising the step of changing the
band select bit to a logic one to shift the RF output spectrum from a
negative side of the RF sinusoidal signal to a positive side of the RF
sinusoidal signal.
29. The method of claim 24, further comprising the steps of coupling the
RF output signal to a matching network; and coupling the output of the
matching network to an antenna.
30. The method of claim 24, further comprising the step of coupling the RF
output signal to an antenna.
31. The method of claim 24, further comprising the step of altering the
frequency of both of the IF I and Q sinusoidal signals in discrete
values.
32. The method of claim 31, further comprising the step of varying the
discrete values in equal frequency steps.
33. A UWB receiver architecture with a first and second RSSI portion to
avoid an interference signal within a multi-band input signal comprising;
at least one integrated substrate; an antenna; an element comprising; a
first mixer coupled to the antenna and at least one of a first quadrature
LO signals; a second mixer coupled to the output of the first mixer and a
second I LO signal; a third mixer coupled to the output of the first
mixer and a second Q LO signal; and an adder/subtractor input coupled to
the output of the second mixer; whereby an output of the third mixer of
the second element is coupled to the input of the adder/subtractor of the
first element; an output of the third mixer of the first element is
coupled to the input of the adder/subtractor of the second element; the
output of the adder/subtractor of the first element is coupled to the
first RSSI portion; the output of the adder/subtractor of the second
element is coupled to the second RSSI portion; whereby the first or
second RSSI portions generates an enable signal if an interference signal
is detected; and the enable signal is coupled to a state machine; whereby
the state machine causes the receiver architecture to hop to a new
channel in the multi-band input signal to avoid the interference signal.
34. The UWB receiver architecture of claim 33, wherein the receiver
architecture resides on the integrated circuit substrate.
35. The UWB receiver architecture of claim 33, wherein the antenna is
formed on first integrated circuit substrate and the remaining receiver
architecture resides on a second integrated circuit substrate.
36. The UWB receiver architecture of claim 33, wherein the state machine
is a DSP, ASIC or FPGA.
37. The UWB receiver architecture of claim 33, further comprising a Low
Noise Amplifier (LNA) that couples the antenna to the first and second
elements.
38. The UWB receiver architecture of claim 33, further comprising a band
select signal with a first and second state; wherein the first state of
the band select signal combines the inputs to the adder/subtractor to
enhance a desired signal and eliminate an image signal; and the second
state of the band select signal combines the inputs to the
adder/subtractor to eliminate a desired signal and enhance an image
signal.
39. The UWB receiver architecture of claim 33, further comprising a Low
Pass Filter and a Programmable Gain Amplifier coupled between each output
of the adder/subtractor and the corresponding RSSI portion.
40. The UWB receiver architecture of claim 33, wherein the second Q LO
signal is in quadrature with the second I LO signal.
41. The UWB receiver architecture of claim 33, wherein the first
quadrature LO and the set of the second I and Q LO signals have discrete
frequency values.
42. The UWB receiver architecture of claim 41, wherein the first
quadrature LO signals are set to a frequency different from the set of
the second I and Q LO signals.
43. The UWB receiver architecture of claim 41, wherein; the first
quadrature LO signals are set to a constant frequency value.
44. The UWB receiver architecture of claim 33, wherein the first and
second RSSI portions each consists of a first and second filters.
45. The UWB receiver architecture of claim 44, further comprising a
comparator which compares the output of the first filter with a reference
signal; wherein a first digital state output of the comparator represents
the presence of an interfering signal; and a second digital state output
of the comparator represents the absence of an interfering signal.
46. The UWB receiver architecture of claim 44, wherein the first and
second filters have non-overlapping frequency characteristics.
47. The UWB receiver architecture of claim 44, wherein the second filter
passes the baseband signal to an Analog to Digital (A/D); whereby the A/D
is coupled to a baseband processing unit for further processing.
48. The UWB receiver architecture of claim 47, wherein the baseband
processing unit is a DSP, ASIC or FPGA.
49. An UWB receiver architecture with an RSSI portion comprising; means
for extracting an RF signal from an antenna; first means for mixing the
RF signal and a first quadrature LO to form an IF signal; second means
for mixing the IF signal and a second quadrature LO to form an I and Q
baseband signals; means for detecting the presence of an interference
signal in the baseband signals using the RSSI portion means; and means
for hopping to a different frequency band to avoid the interference
signal means.
50. The transmitter architecture of claim 49, wherein the first and second
quadrature signals have discrete frequency values.
51. The transmitter architecture of claim 49, wherein the first quadrature
LO is set to a frequency different than that of the second quadrature LO.
52. The transmitter architecture of claim 49, wherein the first quadrature
LO has a frequency that is constant.
53. The UWB receiver architecture of claim 49, wherein the interference
signal is a narrow band signal.
54. The UWB receiver architecture of claim 53, wherein the narrow band
signal is an 802.11 WLAN signal.
55. The UWB receiver architecture of claim 53, wherein the narrow band
signal is a cellular signal.
56. A method of avoiding an interference signal in an UWB receiver
comprising the steps of; using a quadrature RF LO sinusoidal signal to
downconvert a multi-band signal to an in-phase IF signal and a
quadrature-phase IF signal; selecting a quadrature IF LO sinusoidal
signal to further downconvert the in-phase IF signal and quadrature-phase
IF signal to an in-phase zero IF signal and a quadrature-phase zero IF
signal; combining components of the in-phase zero IF signal and the
quadrature-phase zero IF signal using a band select signal to delete an
image band and enhance a desired signal band; applying the desired signal
band of the baseband I signal to a first RSSI portion; applying the
desired signal band of the baseband Q signal to a second RSSI portion;
detecting if the interference signal is present using the first or second
RSSI portions; and hopping to a new channel within the multi-band signal;
thereby avoiding the interference signal in an UWB receiver.
57. The UWB receiver architecture of claim 56, wherein the first and
second RSSI portions each consists of a first and second filters.
58. The UWB receiver architecture of claim 57, wherein the second filter
passes the signal to an Analog to Digital (A/D) for further processing by
a baseband processing unit.
59. The UWB receiver architecture of claim 58, wherein the baseband
processing unit is a DSP, ASIC or FPGA.
60. The UWB receiver architecture of claim 57, wherein the first and
second filters have non-overlapping frequency characteristics.
61. The UWB receiver architecture of claim 57, further comprising the step
of comparing the output of the first filter with a reference signal;
wherein a first digital state output of the comparator represents the
presence of an interfering signal; and a second digital state output of
the comparator represents the absence of an interfering signal.
62. The UWB receiver architecture of claim 61, wherein the digital state
output of the comparator is applied to a state machine; whereby a
decision is made to hop to the new channel.
63. The UWB receiver architecture of claim 62, wherein the state machine
is a DSP, ASIC or FPGA.
64. A UWB receiver architecture with a first and second RSSI portion to
avoid an interference signal comprising; a multi-band signal coupled to a
first and second RF mixers; a RF LO oscillator generating a RF I LO and
RF Q LO quadrature sinusoidial signals; the RF I LO is coupled to the
first RF mixer downconverting the multi-band signal to an IF_I signal;
the RF Q LO is coupled to the second RF mixer downconverting the
multi-band signal to an IF_Q signal; a IF LO oscillator generating a IF I
LO and IF Q LO quadrature sinusoidial signals; 1the IF_I signal is
coupled to a first and second IF mixer downconverting the IF_I signal
into a first and second baseband components; the IF_Q signal is coupled
to a third and fourth IF mixer downconverting the IF_Q signal into a
third and fourth baseband components; a first adder/subtractor controlled
by a band select signal is coupled to the first and third baseband
signals and generates the baseband I signal coupled to the first RSSI
portion; a second adder/subtractor controlled by the band select signal
is coupled to the second and fourth baseband signals and generates the
baseband Q signal coupled to the second RSSI portion; and the band select
signal enhances a desired signal and cancels an image signal; wherein the
first and second RSSI portions can detect an interference signal and
cause the receiver to hop to a different frequency range of the
multi-band signal.
Description
CROSS-REFERENCE TO RELATED APPLICATIONS
[0001] The present application is related to the co-filed U.S. application
entitled "METHOD OF FREQUENCY PLANNING IN AN ULTRA WIDE BAND SYSTEM"
filed on Dec. 29, 2005, which are all invented by at least one common
inventor as well as being assigned to the same entity as the present
application and incorporated herein by reference in their entireties.
BACKGROUND OF THE INVENTION
[0002] Ultrawideband (UWB) wireless technology is a high data rate (480+
Mbps), short range (up to 20 meters), and low power technology that
promises to eliminate confusing cables and wires between interfaces. A de
facto standard has emerged and is known as the MultiBand OFDM Alliance
(MBOA). The FCC (Federal Communication Commission) has allocated an
unlicensed radio spectrum from 3.2 GHz to 10.6 GHz for the MBOA-UWB
technology.
[0003] The full bandwidth of 7.5 GHz is broken up into fourteen multiple
carriers each having a 525 MHz bandwidth and in essence forming a
multi-band system. The need to transfer data in one or more of these
multiple carriers is determined in real time where various channels can
be turned off or on under software control depending on the interference
level of similar systems in the local environment.
[0004] Some typical applications include: video to/from computers and TV,
residential gateways, PDA synchronization, and games to name a few.
Various existing standards may utilize the UWB technology such as HDTV,
SD, Media PC's, and video recorders.
[0005] The MBOA UWB specification has a total bandwidth of 3-10 GHz. As
illustrated in FIG. 1a, the column BAND_ID of table 1-1 shows a total of
14 channels grouped into five band groups. The band groups 1 through 4
have three channels each while band group five has two channels. FIG. 1b
illustrates the spectrum of the 14 channels of the UWB band where the
center frequencies of each channel are identified. For example, spectrum
1-2 corresponds to the BAND_ID channel 7 located within the third Band
Group of table 1-1.
[0006] The MBOA UWB specification requires very fast channel switching
time in 3-10 GHz band. Devices operating in the first four groups require
channel switching less than 9.47 nS.
[0007] An UWB specification is being formed in the 60 GHz range as well.
FIG. 1c illustrates the bandwidth of this spectrum for three regions:
Japan, Europe and the U.S. Note that oxygen can absorb some of the
electromagnetic energy in this range.
[0008] The implementation of UWB RF transceiver imposes several design
constrains in term of frequency planning. They are: the required
switching time, the total number of synthesizers to cover wide frequency
range and the frequency divider operating speed. Since it is very
difficult to implement a synthesizer that can switch up to 1024 MHz in
9.47 nS the synthesizers need to be always enabled so that they generate
a constant frequency without the need to alter the frequencies. This
requires careful frequency planning to minimize the number of
synthesizers and to allow for a feedback divider in the synthesizer at
the lowest possible speed to enable a robust manufacturing yield. The
hopping pattern Time Frequency Code in UWB specifies every band group
needs to hop at least 3 channels while the 9.47 nS comes from the UWB
specifications. The frequency plan can be improved by decreasing the need
for requiring a single synthesizer for each channel. For instance, some
frequencies can be obtained by dividing a higher synthesizer frequency by
a multiple of two several times.
[0009] FIG. 2a illustrates a direct down conversion architecture 2-1. An
antenna 2-2 and matching circuit (not shown) receives the external signal
2-3. A low noise amplifier 2-4 amplifies the received signal and applies
it to the two mixers 2-5 and 2-6. A quadrature oscillator signal
consisting of a I (in-phase) and a Q (quadrature) sinusoidal signals are
applied to the two mixers 2-5 and 2-6 to downconvert the received signal
into an IF_I and an IF_Q component, respectfully. These I and Q
oscillator signals are also know as the local oscillators (LO). Low Pass
Filters (LPF) 2-7 and 2-8 filter out the high frequencies components
before applying the signal to the Programmable Gain Amplifier (PGA) 2-9
and 2-10. The output signals are the baseband I 2-11 and baseband Q 2-12.
[0010] FIG. 2b depicts the fixed local oscillators that can be selected by
a multiplexer or MUX 2-14 to generate the I/Q signals 2-16 which can be
applied, for example, to the mixers 2-5 and 2-6. These synthesizer or
oscillator signals 2-15 or a subset of them can be sourced from an
external lead, ring oscillators, LC tank circuits or Phase Lock Loops
(PLL). The MUX 2-14 selects the I/Q signal that is applied to the down
conversion architecture 2-1. Because these channel frequencies do not
share a common denominator at a reasonable high frequency; up to 14
separate synthesizers may be required. Only one of the 14 LO's 2-15 is
selected by the MUX 2-14 using the channel select signal 2-13.
[0011] Potential issues for the MUX 2-14 shown in FIG. 2b are as follows.
The higher the channel frequency then there is a possibility for more
quadrature mismatch, for instance, the I and Q signals may not be
90.degree. out of phase with each other. In addition, carrier leakage may
occur in the mixers degrading the recovered signal, and finally the
receiver may suffer an output dc offset. Finally, in order to meet the
9.47 nS switching time that was mentioned earlier, the synthesizer
requires that all 14 PLLs are in continuous operation which will cause
high power dissipation levels.
[0012] The process technology and die yield is critical for low cost
consumer product such as UWB devices. At 10 GHz, it may be difficult for
the synthesizer feedback divider to fully function over PVT (Process,
voltage, and Temperature) unless the design uses an advanced technology
process which will increase the cost. Furthermore, forming 14 separate
synthesizers may demand quite large die area increasing the overall cost
of the die.
[0013] A variation of LO generation system 2-17 includes the use of one or
more low speed synthesizers with several single side band (SSB) mixers to
generate the required LO as shown in FIG. 2c. For example, the reference
clock oscillator 2-18 is applied to the PLL 2-19 which generates a 6336
MHz signal on lead 2-20. The divide by 3 2-21 generates a 2112 MHz signal
on lead 2-22. The divide by 2 2-23 generates a 1056 MHz signal on lead
2-24. Assume that the selector block 2-25 passes the signal on lead 2-24
to the SSB 2-26. If the selector block 2-27 passes the signal at the
output of the SSB 2-26, the signal 2-28 will generate a signal frequency
of 5280 MHz on its output lead where this signal needs to be in
quadrature. The signal path from the 6336 MHz LO to the final output
consist of I and Q signals, which require twice the number of dividers
and selectors as shown in FIG. 2c. Note from FIG. 1a, in table 1-1, this
corresponds to the Lower Frequency for BAND_ID channel 5.
[0014] For the circuit illustrated in FIG. 2c, the operating frequency of
the divider is reduced. Each side band mixer will consume inductor area
and power. Due to device mismatch, quadrature mismatch and limited
linearity, this SSB mixer will produce spurs at the transmitter output.
In addition, the receiver will be sensitive to interference at the spur
locations. Moreover, the issues such as the quadrature mismatch, centered
carrier leakage, receiver output dc offset still exists in this method.
[0015] FIG. 3 illustrates a low noise amplifier (LNA) 3-5 feeding a Weaver
architecture 3-1. The external signal 3-2 arrives at the antenna 3-3.
This antenna may exist off-chip or can be integrated on-chip. In
particular, as the frequency of the RF signal increases (60 GHz), the
physical size of the antenna decreases encouraging on-chip formation. A
Band Pass Filter (BPF) 3-4 is used to filter some signal components
before being applied to the LNA 3-5. A first adjustable LO 3-8 is used to
downconvert the signal using the mixers 3-6 and 3-7 into I and Q
components. A BPF 3-9 and 3-10 are used to filter the signal. A second
constant LO 3-11 phase shifts one path by 90.degree. with respect to the
other and is again filtered by BPF's 3-14 and 3-15. The desired signal in
each path to be enhanced by a factor of 2 while the summer 3-16 causes
the image signal to be cancelled after passing through the summer 3-16.
The final signal is available at lead 3-17. The Weaver architecture uses
the energy in the desired signal between the two paths and combines them
together while the image signal is configured to have an opposite
polarity such that the combination eliminates the image components.
BRIEF SUMMARY OF THE INVENTION
[0016] The present invention provides for a simpler technique to insert a
signal into a multi-channel communication system. This technique uses a
modified Weaver architecture in conjunction with adder/subtractor
components in the transmitter to insert a signal into the multi-channel
system. In this architecture, the image is eliminated while the desired
signal is enhanced after passing through this new architecture. The
adder/subtractor components under control of a band select bit
manipulates the upconverted signal twice in the transmitter. The first
situation is after the IF mixers while the second situation occurs after
the RF mixer.
[0017] The Weaver architecture is used to extract the baseband I and Q
signals from the signal content and generate the I and Q baseband
signals. In addition, since the entire signal is captured after the first
LO conversion due to the Weaver architecture, the efficiency of this
architecture is improved. The invention adds an RSSI circuit to the MBOA
receiver to detect whether an 802.11 WLAN signal is interfering with the
desired UWB signal. If so, the system is designed to detect this
interference and jump to a new frequency range to avoid this
interference.
[0018] Thus, this new form of architecture for UWB offers advantages on
several fronts. The selection of the band select bit causes the
upconverted IF baseband I and Q signals to form on either side of an RF
LO signal. Thus, by a simple change a digital bit, one of two RF
bandwidths can be filled. In addition, the signal insertion is enhanced
since a modified Weaver architecture is used. Our invention focuses on
devices that operate over entire UWB band but can be applied for devices
for limited band of operations.
BRIEF DESCRIPTION OF THE SEVERAL VIEWS OF THE DRAWINGS
[0019] FIG. 1a depicts a table indicating the 14 frequencies bands of the
MBOA.
[0020] FIG. 1b illustrates the spectrum of a UWB system from 3 GHz to 10
GHz.
[0021] FIG. 1c illustrates the spectrum of a UWB system at 60 GHz.
[0022] FIG. 2a shows a direct conversion architecture generating baseband
I and Q signals.
[0023] FIG. 2b depicts a MUX selecting one of all available LO
frequencies.
[0024] FIG. 2c illustrates a simplified system that reduces the number of
required LO frequencies.
[0025] FIG. 3 shows a Weaver architecture.
[0026] FIG. 4a shows a first architecture for frequency planning in
accordance with the present invention.
[0027] FIG. 4b shows the frequency spectrum for frequency planning in
accordance with the present invention.
[0028] FIG. 4c shows a second architecture for frequency planning in
accordance with the present invention.
[0029] FIG. 4d shows a third architecture for frequency planning in
accordance with the present invention.
[0030] FIG. 5 depicts fourth architecture for frequency planning in
accordance with the present invention.
[0031] FIG. 6a depicts the signal spectrum, the LO, IF LO, and baseband
signal in accordance with the present invention.
[0032] FIG. 6b lists the frequencies of the RF, LO and IF in accordance
with the present invention.
[0033] FIG. 7a illustrates a prior art Weaver receiver architecture.
[0034] FIG. 7b illustrates receiver architecture for frequency planning in
accordance with the present invention.
[0035] FIG. 7c illustrates an RSSI measurement technique in accordance
with the present invention.
[0036] FIG. 8 illustrates the frequency planning at 60 GHz in accordance
with the present invention.
[0037] FIG. 9 illustrates a receiver architecture at 60 GHz in accordance
with the present invention.
[0038] FIG. 10 illustrates a transmitter architecture at the 3 GHz to 10
GHz range in accordance with the present invention.
[0039] FIG. 11 illustrates a transmitter architecture at the 60 GHz range
in accordance with the present invention.
DETAILED DESCRIPTION OF THE INVENTION
[0040] A simple conceptual diagram 4-1 of one aspect of the invention is
illustrated in FIG. 4a. The antenna and LNA have been described before
and will not be further discussed. A constant LO signal 4-2 is applied to
the RF mixers. Assume that the RF frequency band consists of two bands
4-7 and 4-8, each 2.DELTA. wide, as illustrated in the diagrams 4-6 of
FIG. 4b. The IF LO mixers have an I and Q signal applied at a frequency
of +.DELTA. 4-3. The baseband signal 4-9 is indicated in the lowest
waveform of FIG. 4b. Note by changing the band select signal 4-5 to the
adder/subtractor 4-4, the other band can be extracted. When one band is
extracted, the other band appears as the image and is subtracted out.
This is a Weaver architecture with the following exceptions: the LO to
the RF mixer remains constant and is positioned at the mid-point of the
band spectrum, an IF LO oscillator with the I and Q sinusoid is applied
to 4-3 with a frequency that is half the bandwidth of one of the two
bands 4-7 and 4-8 and creates a zero-IF response, and finally, the band
select signal 4-5 selects which one of the two bands 4-7 and 4-8 is the
image and passes the other band as the extracted signal. The extracted
signal is available at the output of the adder/subtractor 4-4.
[0041] A slight modification of the invention 4-10 is depicted in FIG. 4c.
The RF frequency contains 4 bands each with a bandwidth of 2.DELTA.. The
LO for the RF mixer is positioned between the second and third bands. The
IF mixer uses the switch 4-11 to select either the +.DELTA. 4-12 or the
+3.DELTA. 4-13 and applies this LO to the IF mixer. The band select
determines whether one of the bands in the positive or negative
frequencies is selected. FIG. 4d illustrates a similar architecture 4-15
as in FIG. 4c except that the I and Q sinusoidal frequencies are flipped
to the IF mixers. This results in the generation of a baseband Q signal
4-16 instead of the baseband I signal 4-14.
[0042] Conceptually, the two architectures and frequency plan given in
FIG. 4c and FIG. 4d can be combined into one receiver architecture 5-1 as
shown in FIG. 5 and enhanced. The RF mixers 5-13 and 5-14 can be shared
between the two architectures 4-10 and 4-15. This is a reduction in area
and potentially can result in lower power.
Theory of Operation: RX
[0043] At the antenna input, the desired RF signal along with an image
exists. This is shown in equation 1. Here the .omega..sub.carrier equals
the summation of the two local oscillators or
.omega..sub.LO1+.omega..sub.LO2. In addition, .omega..sub.signal is
equivalent to the baseband signal
-.omega..sub.baseband.RF.sub.in=A.sub.rf
cos{((.omega..sub.carrier+.omega..sub.signal).times.t}+A.sub.image
cos{(.omega..sub.image.times.t} (1)
[0044] The first quadrature RF mixers 5-13 and 5-14 in FIG. 5 translate
the incoming RF signal using equation 1 into an I and Q IF signal
components. Notice that the quadrature RF LO1 oscillators can distinguish
the image signal which is on one side (negative side) of the LO2 from the
desired signal which is on the other side (positive side) of LO2.
Equation 2 and equation 3 reinforce this aspect.IF.sub.1=[A.sub.rf
cos{(.omega..sub.carrier+.omega..sub.signal).times.t}+A.sub.image
cos{.omega..sub.rf.sub.--.sub.image.times.t}]
cos(.omega..sub.LO1.times.t)=1/2[A.sub.rf
cos{(.omega..sub.LO2+.omega..sub.signal).times.t}+A.sub.image
cos{(.omega..sub.LO2-.omega..sub.if.sub.--.sub.image).times.t}]
(2)IF.sub.Q=[A.sub.rf
cos{(.omega..sub.carrier+.omega..sub.signal).times.t}+A.sub.image
cos{.omega..sub.rf.sub.--.sub.image.times.t}]
sin(.omega..sub.LO1.times.t)=1/2[A.sub.rf
sin{(.omega..sub.LO2+.omega..sub.signal).times.t}-A.sub.image
sin{(.omega..sub.LO2-.omega..sub.if.sub.--.sub.image).times.t}] (3)
[0045] The outputs of IF signals are furthered down-converted by the I and
Q LO2 signals in the baseband IQ mixers 5-6 through 5-9 to generate
BB.sub.II, BB.sub.QQ, BB.sub.IQ and BB.sub.QI. Notice that image
component between two corresponding equations have a sign difference
(compare equation 4 and equation 5). This aspect can be used to cancel
the image. In addition, by changing the polarity of the select bit, the
opposite situation occurs. In this case, the image is passed while the
signal is cancelled.BB.sub.II=1/2[A.sub.rf
cos{(.omega..sub.LO2+.omega..sub.signal).times.t}+A.sub.image
cos{(.omega..sub.LO2-.omega..sub.if.sub.--.sub.image).times.t}].times.cos-
(.omega..sub.LO2.times.t)=1/4[A.sub.rf
cos(.omega..sub.signal.times.t)+A.sub.image
cos(.omega..sub.image.times.t)] (4)BB.sub.QQ=1/2[A.sub.rf
sin{(.omega..sub.LO2+.omega..sub.signal).times.t}-A.sub.image
sin{(.omega..sub.LO2-.omega..sub.if.sub.--.sub.image).times.t}].times.sin-
(.omega..sub.LO2.times.t)=1/4[A.sub.rf
cos(.omega..sub.signal.times.t)-A.sub.image
cos(.omega..sub.image.times.t)] (5)BB.sub.IQ=1/2[A.sub.rf
cos{(.omega..sub.LO2+.omega..sub.signal).times.t}+A.sub.image
cos{(.omega..sub.LO2-.omega..sub.if.sub.--.sub.image).times.t}].times.sin-
(.omega..sub.LO2.times.t)=1/4[A.sub.rf
sin(.omega..sub.signal.times.t)+A.sub.image
sin(.omega..sub.image.times.t)] (6)BB.sub.QI=1/2[A.sub.rf
sin{(.omega..sub.LO2+.omega..sub.signal).times.t}-A.sub.image
sin{(.omega..sub.LO2-.omega..sub.if.sub.--.sub.image).times.t}].times.cos-
(.omega..sub.LO2.times.t)=1/4[A.sub.rf
sin(.omega..sub.signal.times.t)-A.sub.image
sin(.omega..sub.image.times.t)] (7)
[0046] The selection of the desired Channel ID is determined by the three
bit signal called Channel Select. One of seven LO2 ranging from 264 MHz
to 3432 MHz is selected and applied to the signal wire 5-5. Each LO2
frequency has an in phase and quadrature phase component.
[0047] The purpose of Band Select bit 5-10 is to select the signal located
either on the positive side or the negative side of the 6864 MHz LO1 that
was applied to the center of the UWB bandwidth. For example, to receive
the signal in the channel located at 7128 MHz, the 264 MHz LO2 is
selected and applied to the IF mixers. Since the LO1 frequency is set at
a constant 6864 MHz value, the image signal is at 6600 MHz. Because of
the Weaver architecture, the image signal is eliminated as indicated in
equation 8 and equation 9 when the band select equals
1.BB.sub.I=BB.sub.II+BB.sub.QQ=1/2[A.sub.rf
cos(.omega..sub.signal.times.t)]
(8)BB.sub.Q=BB.sub.IQ+BB.sub.QI=1/2[A.sub.rf
sin(.omega..sub.signal.times.t)] (9)
[0048] Similar argument can be applied to extracting the signal in located
on the negative side of LO1. The band select bit is set to 0 as shown in
equation 10 and equation 11 when the band select equals
0.BB.sub.I=BB.sub.II-BB.sub.QQ=1/2[A.sub.image
cos(.omega..sub.image.times.t)]
(10)BB.sub.Q=BB.sub.IQ-BB.sub.QI=1/2[A.sub.image
sin(.omega..sub.image.times.t)] (11)
[0049] A Weaver Architecture with variable-zero-IF is provided by the lead
5-5 and is applied to the second set of mixers 5-6 to 5-9. For a
comparison, the traditional Weaver Architecture shown in FIG. 3 selects
the channel or band by using a variable LO 3-8 in the RF mixers 3-6 and
3-7. The second set of IF mixers 3-12 and 3-13 uses the LO 3-11 which has
a fixed IF frequency. The result of this mixings generates a non-zero-IF.
The image signal in the conventional Weaver Architecture is formed only
on one side of the first LO.
[0050] The invention in FIG. 5 exploits the image rejection properties in
the Weaver Architecture even further. In addition, we use variable IF
instead of a fixed IF to perform channel selection. The secondary image
in the Weaver Architecture is also avoided by the use of zero-IF in the
second group of mixers for all channels. Note that the switch 4-11
illustrated earlier in FIG. 4c has been replaced by a MUX controlled by
the channel select signal 5-4. Finally, the band select 5-10 can select
the signal located in either the negative or positive frequencies
surrounding the LO1 frequency.
[0051] This version of the invention only requires six synthesizers 5-3
instead of the fourteen synthesizers mentioned in FIG. 2b to cover entire
UWB band. The six PLL frequency plan consists of one RF PLL and five
IFPLL. The RF PLL generates a quadrature LO at 6864 MHz. The five IFPLL
outputs are at 792, 1320, 1848, 2376 and 2904 MHz.
[0052] The spectrum diagrams for the receiver section 6-1 given in FIG. 6a
describes the operation of the invention, but the same arguments apply to
the transmit section as well. The frequency plan in the receiver section
is now described. The first LO 6-2 for the RF mixers is placed between
the seventh and eighth RF frequency bands as illustrated in FIG. 6a. Note
that this LO generates a constant frequency, unlike the case described in
FIG. 3. This LO at 6864 MHz translates the RF frequencies to IF -3432 to
+3432 MHz as shown in 6-3. FIG. 6b tabulates these frequencies in the
table 6-6. Next the MUX 5-15 in FIG. 5 selects which IF LO frequency 6-4
to select. This downconversion generates the baseband signal 6-5 as the
output. This architecture uses the quadrature mixer in the first
conversion to provide both positive and negative IF signals to the second
mixer. If the positive band is selected, the signal in negative band
becomes image and vice versa.
[0053] Since the IF is symmetrical about DC, only seven LO's are needed to
selects 14 channels in this frequency plan. The unique selection of LO at
6864 MHz has additional advantages. The 6864 Mhz LO can be furthered
divided to obtain the frequencies 264, 528 and 3234 MHz IQ signals. These
signals can be used for the IF LO and as well as 528 MHz sampling clock
for the baseband processor saving three synthesizers. The UWB
specification calls for the sampling clocks to be used from the same
clock.
[0054] One of main advantage of this architecture is that the dividers are
operating at half of maximum RF frequency, thereby, saving power
consumption and complexity. IQ mismatch is reduced by a factor of three
since the maximum IF frequency is 3432 MHz.
[0055] Second, the LO input to the RF mixers does not need multiplexing
which saves power consumption. Consider a direct conversion, for example,
the LO signal would need to be multiplexed and applied to the RF mixers
depending on the channel. The range of the RF LO can extend from 3 to 10
GHz. This would require a large number of RF LO's at high frequencies and
consume significant amounts of power. In present invention, the
multiplexing is only done only at IF mixers which occurs at a lower
frequency range of 264 MHz to 3432 MHz and dissipates much less power.
[0056] Third, all the IF VCO can be implemented with ring oscillators
instead of LC oscillator. The area occupied a LC oscillator is
significantly larger than a ring oscillator. Thereby significant
reduction dies area occupied by LO generation blocks.
[0057] Assuming IQ mismatches mainly comes from LO phases, low IF
frequency has robust IQ accuracy. This translates to better receive and
transmit EVM (Error Vector Magnitude).
[0058] One of the tough specifications of UWB is that center carrier
leakage is in the transmit spectrum mask. It is well known that the
leakage is due to DC offset of the I/Q modulator and leakage through the
LO switches. The second path is frequency dependent. The choice of the
lower IF alleviates this problem.
[0059] Spurious performance or image rejection is only limited by the
first LO IQ accuracy at 6864 MHz, which is easier to achieve than
attempting to perform the first LO I/Q at 10 GHz.
[0060] FIG. 7A illustrates a Weaver receiver architecture that is used in
cellular narrow band systems. This figure has been extracted from a
Hajimiri et. al. patent , U.S. Pat. No. 6,917,815, hereafter called
"Hajimiri". The invention presented in this specification overcomes
several shortcomings pointed out by Hajimiri.
[0061] As indicated in Hajimiri in the second paragraph of column 9; "In
the concurrent downconversion scheme, however, since the unwanted image
signal is one of the two desired signal bands, there is no attenuation of
the image by any of the antenna, the front-end bandpass filter or the
dual-band LNA. Thus, one must rely solely on the image rejection of
Weaver's single sideband downconverter, which is limited by the phase and
amplitude mismatch of the quadrature local oscillators and signal paths,
and can only provide about 20-40 dB attenuation of the unwanted image in
each band. This is clearly insufficient image rejection for the
intermediate frequency signals and thus fails as a solution to the
concurrent dual-band problem."
[0062] Our invention shows how the image rejection issue raised by
Hajimiri is not a problem in the Weaver architecture proposed for the UWB
system that is described in this specification. This basically occurs
because while Hajimiri deals with a narrow band cellular signal, while
the UWB system is a wide band signal. Unlike the UWB system, a typical
GSM cellular system needs to deal with signal levels as low as -106 dBm.
Due to wide bandwidth of UWB systems, the noise floor is -86 dBm without
processing gain.
[0063] Thus, the issues limiting Hajimiri do not have an influence or can
be significantly reduced in the UWB architecture system. A second
important issue is the maximum power levels of the cellular and UWB
systems. The UWB has much lower power levels. A cell phone tower can
transmit as much as 30 dBm while the UWB system transmits only -41
dBm/MHz with peak power of -27 dBm.
[0064] The UWB system is designed for Personal Area Network (PAN)
applications. In such a typical application, there can be several UWB
transmitters within a given PAN area. Consider the case of only two
transmitters. The first transmitter antenna is located 1 meter from the
receiver antenna and acts as an interferer. The second UWB transmitter
antenna is 15 meters away from the receiver antenna and transmits channel
information which is desired to be captured, received and processed by
the receiver.
[0065] The interferer signal sustains a loss while propagating in free
space to the receiver's antenna. This loss can be determined by using the
standard "Friis" equation, which can be used to determine the free space
loss between isotropic radiators and is defined as:Loss (in dB)=[32.44+20
log (dist in km)+20 log (freq in MHz)]dB (12)
[0066] Equation 12 is used to determine the minimum case path loss at two
different frequencies (where k=7 and 1, respectively) at -k frequencies
with regard to the center of the UWB bandwidth spectrum. For the case of
k=-7, the frequency band of 3.4 GHz has a loss of 43 dB after propagating
through a 1 meter distance. If k=-1, the frequency band of 6.6 GHz has a
loss of 49 dB after propagating through a 1 meter distance.
[0067] In present submicron technology, with careful layout and well
characterized foundry device mismatch data, an image rejection 35-45 dB
can be achieved depending on the channel frequency of our IF
architecture. It can be shown that image rejection is function of
frequency. Our unique architecture further relaxes the matching
requirement since our maximum IF is 3432 MHz. This eliminates the need
for the complicated DUAL-BAND FRONT TRANSFER FUNCTION as described in
FIG. 9 of Hajimiri.
[0068] It can be shown that the UWB system architecture presented in this
specification offers several features over the previous prior art. The
first aspect allows robust operation over this range of image rejection
values. In addition, a second aspect does not require the LO frequency of
the RF to IF conversion to have an offset from the mid-point of the
desired signal and the image signal.
[0069] The FCC requires that the UWB transmitters have a maximum power
level of -41 dBm/MHz. The -41 dBm signal is an average power which can
attain a peak power as high as -27 dBm. As long as both the signal and
the image contain UWB signals, an average power of -41 dBm can be assumed
in the following analysis.
[0070] Thus, in the case of the interferer UWB transmitter, the previous
information of the path loss, maximum power level and the image rejection
values can be used to determine the interference signal level of the
image signal.
[0071] Case one: A Nearby UWB Jammer
[0072] For the case of the nearby UWB TX (located at 1 m from the
receiver), the maximum power level is given as -41 dBm [average power].
Use k =-7 and -1, respectively, as before for the image signal band of
6864-k*IF. At the receiver's antenna, this signal will experience a
minimum loss of 43-49 dB, respectively. Since the maximum power level is
-41 dBm, the interference signal level at the antenna when k=-7 is, -41
dBm -43 dB=-84 dBm. Similarly, the interference signal level at the
antenna when k=-1 is, -41 dBm -49 dB=-90 dBm.
[0073] As mentioned earlier, the image rejection can range between 35-45
dB. Thus, when k=-7, the inference signal level of the nearby UWB TX will
be -84 dBm-35 dB=-19 dBm while the interference level will be -129 dBm
for the case of an image rejection of 45 dB. For the case where k=-1, the
inference signal level of the nearby UWB TX will be -90 dBm-35 dB=-125
dBm while the interference level will be -135 dBm for the case of an
image rejection of 45 dB.
[0074] The next important parameter to determine is the thermo noise floor
of a UWB signal which indicates the boundary between a potentially
detectable signal and noise. Since the UWB signal bandwidth is 528 MHz,
the thermo noise floor for the UWB system can be determined by using the
following relationship given in equation 13:Thermo noise floor (dBm)=-174
dBm/Hz+10*log(528*1E6)=-86.7 dBm. (13)
[0075] Thus, the maximum detectable signal level of the UWB signal is
-86.7 dBm. Anything below this value is considered as noise. A UWB
receiver requires 4 dB to 20 dB of SNR depending on the data rate. A
typical receiver has sensitivity threshold set to -86.7 dBm+SNR. As long
as the signal level is less than -82.7 dBm, the packet of information
will not be detected.
[0076] The interference level determined earlier of the jamming UWB signal
ranged from -119 dBm to -135 dBm. This implies that the jamming signal
ranges from 33 dB to 49 dB below the thermo noise level, thus the image
rejection of a nearby UWB jamming signal is not a limiting factor in the
limitation of the system. Therefore, the present invention is not
influenced by a nearby UWB jamming signal and the architecture is a
viable solution to UWB system.
[0077] In addition, assume that the image rejection is increased to 20 dB,
the upper range mentioned by Hajimiri. The jamming signal ranges from 18
dB to 34 dB below the thermo noise level. In some cases, although it is
an extreme example, the UWB system may still operate. Next, a second case
will be considered for a WLAN interferer.
[0078] Case 2: A WLAN interference signal
[0079] In PAN applications, besides a UWB interfering signal, WLAN devices
(e.g., 802.11) can create an undesired interference signals. The WLAN
output power levels can be as high as 20 dBm within a bandwidth of 20
MHz. This high power level will cause the UWB receiver system to fail if
the WLAN transmitter is 1 m away and the WLAN signal falls right on top
of either image or signal channel. The WLAN signal desensitizes the LNA
and mixer stages, which can become fully saturated. Therefore, the UWB
receiver needs to be cleaver enough to avoid the WLAN interference signal
or increase the linearity of the LNA and mixer. Usually, the linearity
can not be achieved without a compromising effect such as designing a
more power dissipative circuit or using more silicon area. Both of these
design issue constraints can be costly. Another approach to avoid a WLAN
interference signal is preferred.
[0080] A Wireless LAN avoidance Scheme
[0081] One possibility is to use a Receiver Signal Strength Indicator
(RSSI) signal having at least one detector connected to the each of the I
and Q IF mixer outputs. An example of an RSSI circuit 7-3 is illustrated
in FIG. 7c. The output of the IF mixer is connected to the lead 7-4. The
first RSSI circuit 7-5 is a low pass filter and has a 1.5 MHz BW. This
filter is used to determine if the WLAN signals are present. At the
beginning of each WLAN signal, there is a 1.5 MHz short pre-ample pilot
signal which has constant amplitude. Under the presence of a strong WLAN
interference signal, this RSSI output signal will be larger than some
predetermined reference signal VREF 7-8. The comparator 7-7 is used to
enable the signal BUSY_CH 7-10. The BUSY_CH can be used to switch the
transceiver to a different frequency band ID. Since there are two RSSI
circuits; one for the I and one for the Q IF mixer outputs, the two
BUSY_CH's signals can be connected to an OR gate to detect if either the
I or Q IF signal senses the WLAN signal.
[0082] If a WLAN RSSI reading indicates the generation of a BUSY_CH signal
then the baseband processor can instruct the UWB receiver system to hop
to a different frequency band to avoid the WLAN interference signal. This
event may cause the loss of a package of information but will allow the
remaining packets to be received. If a packet was lost, then a request
can made to resend this packet. The current UWB specification does not
specify such requirement and may be a useful technique to integrate into
the UWB system specification.
[0083] When a WLAN signal is not detected, the UWB signal has a 4.25 MHz
sub-carrier spacing, therefore, the Band Pass (4-250 MHz BW) filter 7-6
passes the received signal to the A/D 7-9. The output of the A/D 7-11 is
then sent to the baseband processing unit to extract the signal.
[0084] Hajimiri also indicates the following in the fourth paragraph of
column 9. "By offsetting the first local oscillator frequency LO1 from
the midpoint between bands A and B, as shown in the figure, applying the
Weaver image rejection technique now not only does not suffer from the
aforementioned drawbacks, but actually significantly improves the image
rejection. The key to this solution is to offset the LO1 frequency of the
first stage of the image-rejection architecture from the midpoint of the
two bands of interest in such a way that the image, fIA, of the first
band, fA, falls at the middle attentuation region of the front-end
subsystem transfer function. Similarly, the image of the second, upper
desired band, fB, falls at outside the pass-band of the front-end at fIB
and will also be attenuated."
[0085] The invention given in this specification for the UWB system has
demonstrated that the image signal is not necessarily a critical concern.
Because of this issue, the UWB architecture does not need to offset the
Local Oscillator (LO) as Hajimiri is required to do. Furthermore,
although the range of 35 to 45 dB image rejection has been shown to be
achievable, a possibility exists for certain situations for the UWB
system to operate with much less image rejection.
[0086] The numbers for the frequency plan using the Weaver architecture
for the 60 GHz UWB system is provided the table 8-1 in FIG. 8. The CH
number is given in the first column, while the center frequency is shown
next. The next two columns indicate the LO1 and LO2 frequencies. Finally,
the last column provides the band select bit value. Each channel has a
bandwidth of 1 GHz.
[0087] The same concepts can be applied to cover the architecture 7-1 in
FIG. 7b where only 12 bands are used. In this case, only five
synthesizers are needed. RF LO 7-2 can be at 6336 MHz and two IF 264 MHz,
792 MHz IQ signals can be derived from 6336 MHz LO. Otherwise, this
architecture operates very similar to the one given in FIG. 5.
[0088] The architecture and frequency plan for the 60 GHz receiver is
illustrated in FIG. 9. Both receivers shown in FIG. 7b and FIG. 9 have a
similar architecture. The primary difference is that the RF frequency
range of FIG. 9 extents to 60 GHz. Thus, the architecture given in FIG. 9
needs no further description.
[0089] Theory of Operation: TX
[0090] The architecture and frequency plan for the MBOA transmitter
section 10-1 is shown in FIG. 10. The baseband I and Q signals (10-2 and
10-3) are applied to the PGA which is then Low Pass Filtered (LPF). The
first four quadrature IF mixers translate the incoming baseband signals
to an IF frequency. Depending on the channel select value 10-5, the IF LO
can be selected from 264 MHz to 3432 MHz. The output IF signals are in
phase and quadrature form, and are up-converted by the 6864 IQ LO signal
in the RF mixer. Since each RF mixer generates the desired and image
signal, the I/Q signals can be used to cancel the image. For example, at
the output of RF mixer, the signal can be at 6864+/-k*IF, where k is the
selected channel. Similar to the receiver, if the positive band is
selected, the signal in negative band becomes image and vice versa. A
single band select signal selects k*IF and determines if the signal
resides in the positive or negative range. For example, if we want to
transmit the signal at +3432 then the IF needs to be 3432 MHz and the
band select bit has to be negative. In addition, if the band select bit
selects the positive polarity then, the transmitter output is 6864+IF.
[0091] The frequency translation of the baseband signal .omega..sub.signal
in the transmitter section is described. The first four quadrature mixers
translate the incoming baseband signal to IF frequency using an LO2 IF
carrier selected by the three bit channel select control 10-5. The output
of the up converted IF signals are in phase and quadrature form and are
summed together using the band select signal 10-4. The IF signal is
further upconverted to RF frequencies by the LO1. If band_select=1, the
higher band is selected as indicated in equation 14 and equation 15. The
purpose of band select bit is to distinguish whether the transmitter
output is selected from the positive or negative side of the constant
6864 MHz LO1 clock frequency. For example, if the transmitter generates a
channel at 7128 MHz, a 264 MHz IF is selected. At the IF output, the
incoming baseband signal needs to be on the POSITIVE side of LO2 signal.
This operation is accomplished as indicated in equation 14 and equation
15.IF.sub.--I=cos(.omega..sub.LO2.times.t).times.cos(.omega..sub.signal.t-
imes.t)-sin(.omega..sub.LO2.times.t).times.sin(.omega..sub.signal.times.t)-
=cos{(.omega..sub.LO2+.omega..sub.signal).times.t}
(14)IF.sub.--Q=sin(.omega..sub.LO2.times.t).times.cos(.omega..sub.signal.-
times.t)+cos(.omega..sub.LO2.times.t).times.sin(.omega..sub.signal.times.t-
)=sin{(.omega..sub.LO2+.omega..sub.signal).times.t} (15)
[0092] Since each mixer generates LO+IF and LO-IF or signal and image, the
image portion of this signal can be subtracted out. These two signals
IF_I and IF_Q are then up-converted by a 6864 MHz IQ LO1 oscillator
signal in the RF mixer. The signal at the antenna is given in equation
16.RF_OUT=cos(.omega..sub.LO1.times.t).times.cos{(.omega..sub.LO2+.omega.-
.sub.signal).times.t}-sin(.omega..sub.LO1.times.t).times.sin{(.omega..sub.-
LO2+.omega..sub.signal).times.t}=cos{(.omega..sub.LO1+.omega..sub.LO2+.ome-
ga..sub.signal).times.t} (16)
[0093] If Band_select=0 the lower band is selected as indicated in
equation 17 and equation 18. Similarly, if we want to generate 6600 MHz
at channel, we will choose an IF of 264 MHz. At the output of the IF
mixers, the incoming baseband signal is selected to be on the negative
side of LO2 signal. This operation is accomplished using equation 17 and
equation 18. The IF signal is further up converted to RF frequency by
LO1. Since each mixer generates LO-IF and LO+IF or signal and image, the
image potion is subtracted. This process is done by use of quadrature LO1
signal in as indicated in equation
19.IF.sub.--I=cos(.omega..sub.LO2.times.t).times.cos(.omega..sub.signal.t-
imes.t)+sin(.omega..sub.LO2.times.t).times.sin(.omega..sub.signal.times.t)-
=cos{(.omega..sub.LO2-.omega..sub.signal).times.t}
(17)IF.sub.--Q=sin(.omega..sub.LO2.times.t).times.cos(.omega..sub.signal.-
times.t)-cos(.omega..sub.LO2.times.t).times.sin(.omega..sub.signal.times.t-
)=sin{(.omega..sub.LO2-.omega..sub.signal).times.t}
(18)RF_OUT=cos(.omega..sub.LO1.times.t).times.cos{(.omega..sub.LO2+.omega-
..sub.signal).times.t}+sin(.omega..sub.LO1.times.t).times.sin{(.omega..sub-
.LO2-.omega..sub.signal).times.t}=cos{(.omega..sub.LO1-.omega..sub.LO2+.om-
ega..sub.signal).times.t} (19)
[0094] The architecture and frequency plan for the 60 GHz transmitter
section 11-1 is shown in FIG. 11. Note that the architecture is very
similar to the architecture given in FIG. 10. Thus the detailed
description is not required. Again, the primary difference is that the
frequencies in FIG. 11 have been increased to the 60 GHz range.
[0095] Finally, it is understood that the above descriptions are only
illustrative of the principles of the current invention. In accordance
with these principles, those skilled in the art may devise numerous
modifications without departing from the spirit and scope of the
invention. For example, the UWB specification calls for the sampling
clocks to be used from the same clock but this general technique can be
used with sampling clocks from other sources. The reference clocks can be
obtained from external sources off chip, LC tank circuits or PLL's. The
actual choice of the clock source will depend on a number of issues,
including, area availability, and ease of use. The technology to form the
circuits can be formed using the MOS or BJT technologies, for example. In
addition, the RSSI circuit can be incorporated into all of the receivers
previously described. Also, the matching network associated with the
antenna may be eliminated in certain cases. Finally, the BPF, LPF and
amplifiers (although they may not be shown specifically) can be
incorporated into the design by those skilled in the art.
* * * * *
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